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19-2373; Rev 0; 4/02
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller
General Description
The MAX1953/MAX1954/MAX1957 is a family of versatile, economical, synchronous current-mode, pulse-width modulation (PWM) buck controllers. These step-down controllers are targeted for applications where cost and size are critical. The MAX1953 operates at a fixed 1MHz switching frequency, thus significantly reducing external component size and cost. Additionally, excellent transient response is obtained using less output capacitance. The MAX1953 operates from low 3V to 5.5V input voltage and can supply up to 10A of output current. Selectable current limit is provided to tailor to the external MOSFETs' on-resistance for optimum cost and performance. The output voltage is adjustable from 0.8V to 0.86VIN. With the MAX1954, the drain-voltage range on the highside FET is 3V to 13.2V and is independent of the supply voltage. It operates at a fixed 300kHz switching frequency and can be used to provide up to 25A of output current with high efficiency. The output voltage is adjustable from 0.8V to 0.86VHSD. The MAX1957 features a tracking output voltage range of 0.4V to 0.86VIN and is capable of sourcing or sinking current for applications such as DDR bus termination and PowerPCTM/ASIC/DSP core supplies. The MAX1957 operates from a 3V to 5.5V input voltage and at a fixed 300kHz switching frequency. The MAX1953/MAX1954/MAX1957 provide a COMP pin that can be pulled low to shut down the converter in addition to providing compensation to the error amplifier. An input undervoltage lockout (ULVO) is provided to ensure proper operation under power-sag conditions to prevent the external power MOSFETs from overheating. Internal digital soft-start is included to reduce inrush current. The MAX1953/MAX1954/MAX1957 are available in tiny 10-pin MAX packages. o Low-Cost Current-Mode Controllers o Fixed-Frequency PWM o MAX1953 1MHz Switching Frequency Small Component Size, Low Cost Adjustable Current Limit o MAX1954 3V to 13.2V Input Voltage 25A Output Current Capability 93% Efficiency 300kHz Switching Frequency o MAX1957 Tracking 0.4V to 0.86VIN Output Voltage Range Sinking and Sourcing Capability of 3A o Shutdown Feature o All N-Channel MOSFET Design for Low Cost o No Current-Sense Resistor Needed o Internal Digital Soft-Start o Thermal Overload Protection o Small 10-Pin MAX Package
Features
MAX1953/MAX1954/MAX1957
Ordering Information
PART MAX1953EUB MAX1954EUB MAX1957EUB TEMP RANGE -40C to +85C -40C to +85C -40C to +85C PIN-PACKAGE 10 MAX 10 MAX 10 MAX
Pin Configurations
TOP VIEW
Applications
Printers and Scanners Graphic Cards and Video Cards PCs and Servers Microprocessor Core Supply Low-Voltage Distributed Power Telecommunications and Networking
ILIM 1 COMP FB GND IN 2 3 4 5
10 BST 9 LX DH PGND DL
MAX1953EUB
8 7 6
MAX Patent Pending PowerPC is a trademark of Motorola, Inc.
Pin Configurations continued at end of data sheet.
________________________________________________________________ Maxim Integrated Products
1
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
ABSOLUTE MAXIMUM RATINGS
IN, FB to GND...........................................................-0.3V to +6V LX to BST..................................................................-6V to +0.3V BST to GND ............................................................-0.3V to +20V DH to LX ....................................................-0.3V to (VBST + 0.3V) DL, COMP to GND.......................................-0.3V to (VIN + 0.3V) HSD, ILIM, REFIN to GND ........................................-0.3V to 14V PGND to GND .......................................................-0.3V to +0.3V IDH, IDL ................................................................100mA (RMS) Continuous Power Dissipation (TA = +70C) (derate 5.6mW/C above +70C) ..................................444mW Operating Temperature Range ...........................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = 5V, VBST - VLX = 5V, TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C.) (Note 1)
PARAMETER Operating Input Voltage Range HSD Voltage Range Quiescent Supply Current MAX1954 only (Note 2) VFB = 1.5V, no switching VIN = VBST = 5.5V, VHSD = 13.2V, COMP = GND Rising and falling VIN, 3% hysteresis 2.50 0.8 CONDITIONS MIN 3.0 3.0 1 220 220 2.78 TYP MAX 5.5 13.2 2 350 350 2.95 0.86 x VIN 0.8 0.8 VREFIN 110 5 5 -0.1 MAX1957 only ILIM = GND (MAX1953 only) VILIM = VIN or ILIM = open (MAX1953 only) MAX1954/MAX1957 -0.1 5.67 3.15 6.3 3.5 0.812 0.812 VREFIN + 8mV 160 500 500 1.5 1.5 6.93 3.85 S nA nA V V V/V V/V V UNITS V V mA A A V V
Standby Supply Current (MAX1953/ MAX1957) VIN = VBST = 5.5V, COMP = GND Standby Supply Current (MAX1954) Undervoltage Lockout Trip Level Output Voltage Adjust Range (VOUT) ERROR AMPLIFIER TA = 0C to +85C (MAX1953/MAX1954) FB Regulation Voltage TA = -40C to +85C (MAX1953/MAX1954) MAX1957 only Transconductance FB Input Leakage Current REFIN Input Bias Current FB Input Common-Mode Range REFIN Input Common-Mode Range Current-Sense Amplifier Voltage Gain Low Current-Sense Amplifier Voltage Gain VFB = 0.9V VREFIN = 0.8V, MAX1957 only 0.788 0.776 VREFIN - 8mV 70
2
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 5V, VBST - VLX = 5V, TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C.) (Note 1)
PARAMETER ILIM Input Impedance MAX1953 only VPGND - VLX, ILIM = GND (MAX1953 only) Current-Limit Threshold VPGND - VLX, ILIM = open (MAX1953 only) VPGND - VLX, ILIM = IN (MAX1953 only) VPGND - VLX (MAX1954/MAX1957 only) OSCILLATOR Switching Frequency Maximum Duty Cycle Minimum Duty Cycle SOFT-START Soft-Start Period FET DRIVERS DH On-Resistance, High State DH On-Resistance, Low State DL On-Resistance, High State DL On-Resistance, Low State LX, BST Leakage Current LX, BST, HSD Leakage Current THERMAL PROTECTION Thermal Shutdown Thermal Shutdown Hysteresis SHUTDOWN CONTROL COMP Logic Level Low COMP Logic Level High COMP Pullup Current 3V < VIN < 5.5V 3V < VIN < 5.5V 0.8 100 0.25 V V A Rising temperature 160 15 C C VBST = 10.5V, VLX = VIN = 5.5V, MAX1953/MAX1957 VBST = 18.7V, VLX = 13.2V, VIN = 5.5V VHSD = 13.2V (MAX1954 only) 2 1.5 2 0.8 3 3 3 2 20 30 A A MAX1953 MAX1954/MAX1957 4 3.4 ms MAX1953 MAX1954/MAX1957 Measured at DH MAX1953, measured at DH MAX1954/MAX1957, measured at DH 0.8 240 86 1 300 89 15 4.5 1.2 360 96 18 5.5 MHz kHz % % CONDITIONS MIN 50 85 190 290 190 TYP 125 105 210 320 210 MAX 200 125 235 350 235 mV UNITS k
MAX1953/MAX1954/MAX1957
Note 1: Specifications to -40C are guaranteed by design and not production tested. Note 2: HSD and IN are externally connected for applications where VHSD < 5.5V.
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3
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
Typical Operating Characteristics
(TA = +25C, unless otherwise noted.)
MAX1953 EFFICIENCY vs. LOAD CURRENT
95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 0.1 1 LOAD CURRENT (A) VOUT = 2.5V CIRCUIT OF FIGURE 1 10 VIN = 5V VIN = 3.3V
MAX1953 toc01
MAX1954 EFFICIENCY vs. LOAD CURRENT
MAX1953 toc02
MAX1957 EFFICIENCY vs. LOAD CURRENT
MAX1953 toc03
100
100 VOUT = 2.5V 90 EFFICIENCY (%) 80 70 60 50 40 0.1 1 LOAD CURRENT (A) VOUT = 1.7V
100 90 EFFICIENCY (%) 80 70 60 50 40 0.1 1 LOAD CURRENT (A)
VOUT = 1.25V
VIN = 5V CIRCUIT OF FIGURE 2 10
VIN = 5V CIRCUIT OF FIGURE 3 10
MAX1954 EFFICIENCY vs. LOAD CURRENT
95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 0 5 10 VIN = 12V CIRCUIT OF FIGURE 4 15 20 25 VOUT = 1.8V
MAX1953 toc04
MAX1953 OUTPUT VOLTAGE vs. LOAD CURRENT
MAX1953 toc05
100
2.60
OUTPUT VOLTAGE (V)
2.55 VIN = 5V 2.50 VIN = 3.3V 2.45
CIRCUIT OF FIGURE 1 2.40 0 0.5 1.0 1.5 2.0 2.5 3.0 LOAD CURRENT (A)
LOAD CURRENT (A)
MAX1954 OUTPUT VOLTAGE vs. LOAD CURRENT
MAX1953 toc06
MAX1954 OUTPUT VOLTAGE vs. LOAD CURRENT
MAX1953 toc07
2.60
1.80
2.55 OUTPUT VOLTAGE (V)
1.75 OUTPUT VOLTAGE (V)
2.50 VHSD = VIN = 5V 2.45
1.70 VHSD = VIN = 5V
1.65
2.40 CIRCUIT OF FIGURE 2 2.35 0 1 2 3 4 5 6 LOAD CURRENT (A)
1.60 CIRCUIT OF FIGURE 2 1.55 0 1 2 3 4 5 6 LOAD CURRENT (A)
4
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
Typical Operating Characteristics (continued)
(TA = +25C, unless otherwise noted.)
MAX1957 OUTPUT VOLTAGE vs. LOAD CURRENT
MAX1953 toc08
MAX1953 OUTPUT VOLTAGE vs. INPUT VOLTAGE
MAX1953 toc09
MAX1954 OUTPUT VOLTAGE vs. INPUT VOLTAGE
MAX1953 toc10
1.35
2.60
1.76 1.74 OUTPUT VOLTAGE (V) 1.72 1.70 1.68 1.66 ILOAD = 5A
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
1.30
2.55 ILOAD = 3A 2.50 ILOAD = 0 2.45
ILOAD = 0
1.25 VIN = 5V 1.20
1.15 -3 -2 -1 0
CIRCUIT OF FIGURE 3 1 2 3
CIRCUIT OF FIGURE 1 2.40 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V) 1.64 3.0 3.5 4.0
CIRCUIT OF FIGURE 2 4.5 5.0 5.5
LOAD CURRENT (A)
INPUT VOLTAGE (V)
MAX1954 OUTPUT VOLTAGE vs. INPUT VOLTAGE
MAX1953 toc11
MAX1957 OUTPUT VOLTAGE vs. INPUT VOLTAGE
MAX1953 toc12
2.52 2.51 OUTPUT VOLTAGE (V) ILOAD = 0 2.50 2.49 2.48 2.47 CIRCUIT OF FIGURE 2 2.46 3.0 3.5 4.0 4.5 5.0
1.29
1.27 OUTPUT VOLTAGE (V)
ILOAD = 0
1.25
ILOAD = 5A
1.23
ILOAD = 3A
1.21 CIRCUIT OF FIGURE 3 1.19 5.5 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V) INPUT VOLTAGE (V)
MAX1953 FREQUENCY vs. INPUT VOLTAGE
MAX1953 toc13
MAX1954/MAX1957 FREQUENCY vs. INPUT VOLTAGE
315 310 FREQUENCY (kHz) 305 300 295 290 285 TA = +85C TA = +25C TA = -40C VOUT = 1.25V
MAX1953 toc14
320
1.06 1.04
VOUT = 2.5V
FREQUENCY (MHz)
TA = -40C 1.02 1.00 0.98 0.96 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V) TA = +85C TA = +25C
280 275 270 3.0 3.5
4.0
4.5
5.0
5.5
INPUT VOLTAGE (V)
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5
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
Typical Operating Characteristics (continued)
(TA = +25C, unless otherwise noted.)
MAX1953 LOAD TRANSIENT
MAX1953 toc15
MAX1954 LOAD TRANSIENT
MAX1953 toc16
VOUT AC-COUPLED
100mV/div VOUT AC-COUPLED 100mV/div
3A ILOAD CIRCUIT OF FIGURE 1 400s/div 400s/div 1.5A ILOAD 5A 2.5A
MAX1957 LOAD TRANSIENT
MAX1953 toc17
MAX1953 NO-LOAD SWITCHING WAVEFORMS
MAX1953 toc18
VOUT AC-COUPLED
50mV/div
ILX
2A/div
LX 3A
5V/div
DL
5V/div
ILOAD -3A 400s/div
DH
5V/div
2s/div
6
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
Typical Operating Characteristics (continued)
(TA = +25C, unless otherwise noted.)
MAX1953 FULL-LOAD SWITCHING WAVEFORMS
MAX1953 toc19
MAX1953 SHORT-CIRCUIT SWITCHING WAVEFORMS
MAX1953 toc20
MAX1954/MAX1957 NO-LOAD SWITCHING WAVEFORMS
MAX1953 toc21
ILX
2A/div
ILX LX
5A/div 5V/div
ILX
2A/div
LX DL
5V/div DL 5V/div 5V/div
LX
10V/div
DL DH 5V/div DH 2s/div 4s/div
5V/div
DH
5V/div
10V/div
2s/div
MAX1954/MAX1957 FULL-LOAD SWITCHING WAVEFORMS
MAX1953 toc22
MAX1954/MAX1957 SHORT-CIRCUIT SWITCHING WAVEFORMS
MAX1953 toc23
ILX
2A/div ILX 5A/div 10V/div
LX
10V/div
LX
DL
5V/div
DL
5V/div
DH 4s/div
10V/div
DH 4s/div
10V/div
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7
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
Pin Description
PIN MAX1953 MAX1954 MAX1957 NAME FUNCTION ILIM Sets the Current-Limit Threshold for the Low-Side N-Channel MOSFET, as well as the Current-Sense Amplifier Gain. Connect to IN for 320mV, leave floating for 210mV, or connect to GND for 105mV current-limit threshold. HSD Senses the Voltage at the Drain of the High-Side N-Channel MOSFET. Connect to the high-side MOSFET drain using a Kelvin connection. REFIN Sets the FB Regulation Voltage. Drive REFIN with the desired FB regulation voltage using an external resistor-divider. Bypass to GND with a 0.1F capacitor. Compensation and Shutdown Control Pin. Connect an RC network to compensate control loop. Drive to GND to shut down the IC. Feedback Input. Regulates at VFB = 0.8V (MAX1953/MAX1954) or REFIN (MAX1957). Connect FB to a resistor-divider to set the output voltage (MAX1953/MAX1954). Connect to output through a decoupling resistor (MAX1957). Ground Input Voltage (3V to 5.5V). Provides power for the IC. For the MAX1953/MAX1957, IN serves as the current-sense input for the highside MOSFET. Connect to the drain of the high-side MOSFET (MAX1953/MAX1957). Bypass IN to GND close to the IC with a 0.22F (MAX1954) capacitor. Bypass IN to GND close to the IC with 10F and 4.7F in parallel (MAX1953/MAX1957) capacitors. Use ceramic capacitors. Low-Side Gate-Drive Output. Drives the synchronous-rectifier MOSFET. Swings from PGND to VIN. Power Ground. Connect to source of the synchronous rectifier close to the IC. High-Side Gate-Drive Output. Drives the high-side MOSFET. DH is a floating driver output that swings from VLX to VBST. Master Controller Current-Sense Input. Connect LX to the junction of the MOSFETs and inductor. LX is the reference point for the current limit. Boost Capacitor Connection for High-Side Gate Driver. Connect a 0.1F ceramic capacitor from BST to LX and a Schottky diode to IN.
1
--
--
ILIM
--
1
--
HSD
--
--
1
REFIN
2
2
2
COMP
3
3
3
FB
4
4
4
GND
5
5
5
IN
6 7 8
6 7 8
6 7 8
DL PGND DH
9
9
9
LX
10
10
10
BST
8
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller
Functional Diagram
IN
MAX1953/MAX1954/MAX1957
THERMAL LIMIT
UVLO
MAX1953 MAX1954 MAX1957
SHUTDOWN COMPARATOR
SLOPE COMPENSATION
HSD (MAX1954 ONLY)
0.5V COMP ERROR AMPLIFIER FB PWM CONTROL CIRCUITRY CURRENTSENSE CIRCUITRY
BST
DH
GND
LX IN DL
REFIN (MAX1957 ONLY)
REFERENCE AND SOFT-START DAC SHORT-CIRCUIT CURRENT-LIMIT CIRCUITRY CURRENT-LIMIT COMPARATOR
PGND
CLOCK
ILIM (MAX1953 ONLY)
Typical Operating Circuit
INPUT 3V TO 5.5V
ILIM
IN
BST DH
MAX1953
COMP
LX DL PGND
OUTPUT 0.8V TO 0.86VIN
GND
FB
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9
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
Detailed Description
The MAX1953/MAX1954/MAX1957 are single-output, fixed-frequency, current-mode, step-down, PWM, DCDC converter controllers. The MAX1953 switches at 1MHz, allowing the use of small external components for small applications. Table 1 lists suggested components. The MAX1954 switches at 300kHz for higher efficiency and operates from a wider range of input voltages. Figure 1 is the MAX1953 typical application circuit. The MAX1953/MAX1954/MAX1957 are designed to drive a pair of external N-channel power MOSFETs in a synchronous buck topology to improve efficiency and cost compared with a P-channel power MOSFET topology. The on-resistance of the low-side MOSFET is used for short-circuit current-limit sensing, while the high-side MOSFET on-resistance is used for current-mode feedback and current-limit sensing, thus eliminating the need for current-sense resistors. The MAX1953 has three selectable short-circuit current-limit thresholds: 105mV, 210mV, and 320mV. The MAX1954 and MAX1957 have 210mV fixed short-circuit current-limit thresholds. The MAX1953/MAX1954/MAX1957 accept input voltages from 3V to 5.5V. The MAX1954 is configured with a high-side drain input (HSD) allowing an extended input voltage range of 3V to 13.2V that is independent of the input supply (Figure 2). The MAX1957 is tailored for tracking output voltage applications such as DDR bus termination supplies, referred to as VTT. It utilizes a resistor-divider network connected to REFIN to keep the 1/2 ratio tracking between VTT and VDDQ (Figure 3). The MAX1957 can source and sink up to 3A. Figure 4 shows the MAX1954 20A circuit.
DC-DC Converter Control Architecture
The MAX1953/MAX1954/MAX1957 step-down converters use a PWM, current-mode control scheme. An internal transconductance amplifier establishes an integrated error voltage. The heart of the PWM controller is an openloop comparator that compares the integrated voltagefeedback signal against the amplified current-sense signal plus the slope compensation ramp, which are summed into the main PWM comparator to preserve inner-loop stability and eliminate inductor staircasing. At each rising edge of the internal clock, the high-side MOSFET turns on until the PWM comparator trips or the maximum duty cycle is reached. During this on-time, current ramps up through the inductor, storing energy in a magnetic field and sourcing current to the output. The current-mode feedback system regulates the peak inductor current as a function of the output voltage error signal. The circuit acts as a switch-mode transconductance amplifier and pushes the output LC filter pole normally found in a voltage-mode PWM to a higher frequency. During the second half of the cycle, the high-side MOSFET turns off and the low-side MOSFET turns on. The inductor releases the stored energy as the current ramps down, providing current to the output. The output capacitor stores charge when the inductor current exceeds the required load current and discharges when the inductor current is lower, smoothing the voltage across the load. Under overload conditions, when the inductor current exceeds the selected current-limit (see the Current Limit Circuit section), the high-side MOSFET is not turned on at the rising clock edge and the low-side MOSFET remains on to let the inductor current ramp down. The MAX1953/MAX1954/MAX1957 operate in a forcedPWM mode. As a result, the controller maintains a constant switching frequency, regardless of load, to allow for easier postfiltering of the switching noise.
Table 1. Suggested Components
DESIGNATION C1 MAX1953 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 0.1F, 50V X7R CER Taiyo Yuden UMK107BJ104KA 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG MAX1954 0.22F, 10V X7R CER Kemet C0603C224M8RAC 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 0.1F, 50V X7R CER Taiyo Yuden UMK107BJ104KA MAX1957 3 x 22F, 6.3V X5R CER Taiyo Yuden JMK316BJ226ML 0.1F, 50V X7R CER Taiyo Yuden UMK107BJ104KA 270F, 2V SP Polymer Panasonic EEFUEOD271R 20A CIRCUIT 0.22F, 10V X7R CER Kemet C0603C224M8RAC 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG
C2
C3
10
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
Table 1. Suggested Components (continued)
DESIGNATION C4 MAX1953 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 4.7F, 6.3V X5R CER Taiyo Yuden JMK212BJ475MG 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG -- MAX1954 180F, 4V SP Polymer Panasonic EEFUEOG181R -- MAX1957 270F, 2V SP Polymer Panasonic EEFUEOD271R 270F, 2V SP Polymer Panasonic EEFUEOD271R 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 4.7F, 6.3V X5R CER Taiyo Yuden JMK212BJ475MG 0.1F, 50V X7R CER Taiyo Yuden UMK107BJ104KA -- 1500pF, 50V X7R CER Murata GRM39X7R152K50 470pF, 50V X7R CER Murata GRM39X7R471K50 68pF, 50V COG CER Murata GRM39COG680J50 Schottky diode Central Semiconductor CMPSH1-4 2.7H 6.6A Coilcraft DO3316-272HC Dual MOSFET 20V Fairchild FDS6898A -- 2k 1% 2k 1% 10k 5% 33k 5% 62k 5% 51.1k 5% 270k 5% 20A CIRCUIT 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 10F, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 0.1F, 50V X7R CER Taiyo Yuden UMK107BJ104KA 270F, 2V SP polymer Panasonic EEFUEOD271R 270F, 2V SP polymer Panasonic EEFUEOD271R -- 560pF, 10V X7R CER Kemet C0402C561M8RAC 15pF, 10V C0G CER Kemet C0402C150K8GAC Schottky diode Central Semiconductor CMPSH1-4 0.8H 27.5A Sumida CEP125U-0R8 N-channel 30V International Rectifier IRF7811W N-channel 30V Siliconix Si4842DY 10k 1% 8.06k 1%
C5
C6
--
C7
--
C8
--
--
C9-C13
--
--
C14
-- 270pF, 10V X7R CER Kemet C0402C271M8RAC -- Schottky diode Central Semiconductor CMPSH1-4 1H 3.6A Toko 817FY-1R0M Dual MOSFET 20V 5A Fairchild FDS6898A -- 16.9k 1% 8.06k 1% 9.09k 1% 8.06k 1%
-- 1000pF, 10V X7R CER Kemet C0402C102M8RAC 47pF, 10V C0G CER Kemet C0402C470K8GAC Schottky diode Central Semiconductor CMPSH1-4 2.7H 6.6A Coilcraft DO3316-272HC Dual MOSFET 20V Fairchild FDS6890A --
CC
Cf
D1
L1
N1-N2
N3-N4 R1 R2 R3 RC
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11
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
VIN 3V TO 5.5V C1 10F C6 10F C5 4.7F ILIM IN BST DH RC 33k CC 270pF D1 N1
C2 0.1F
L1 1H
VOUT 2.5V AT 3A C3 10F C4 10F
MAX1953
COMP
LX DL PGND R2 8.06 R1 16.9k
GND
FB
Figure 1. Typical Application Circuit for the MAX1953
VIN 3V TO 5.5V C1 0.22F VHSD 5.5V TO 13.2V D1 IN HSD RC 62k CC 1000pF Cf 47pF BST DH N1
C2 10F
C3 0.1F
L1 2.7H
VOUT 1.7V AT 3A C4 180F
MAX1954
COMP
LX DL PGND R2 8.06k R1 9.09k
GND
FB
Figure 2. Typical Application Circuit for the MAX1954
12
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
VIN 3V TO 5.5V C1 3 22F RC 51.1k COMP VDDQ R1 2k CC 470pF Cf 68pF C6 10F C7 4.7F IN BST DH D1 N1 L1 2.7H C14 1500pF R3 10k
C2 0.1F
VTT = 1/2 VDDQ C3 270F C4 270F C5 270F
MAX1957
REFIN C8 0.1F
LX DL PGND
R2 2k
GND
FB
Figure 3. Typical Application Circuit for the MAX1957
VHSD 10.8V TO 13.2V VIN 3V TO 5.5V D1 C1 0.22F HSD IN BST DH N1 C7 0.1F N3 N4 R1 10k C8 270F C9 270F C10 270F C11 270F C12 270F C13 270F N2 L1 0.8H VOUT 1.8V AT 20A C2 10F C3 10F C4 10F C5 10F C6 10F
RC 270k CC 560pF Cf 15pF
MAX1954
COMP
LX DL PGND
GND
FB
R2 8.06k
Figure 4. 20A Circuit
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13
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
Current-Sense Amplifier
The MAX1953/MAX1954/MAX1957s' current-sense circuit amplifies (AV = 3.5 typ) the current-sense voltage (the high-side MOSFET's on-resistance (RDS(ON)) multiplied by the inductor current). This amplified currentsense signal and the internal-slope compensation signal are summed (VSUM) together and fed into the PWM comparator's inverting input. The PWM comparator shuts off the high-side MOSFET when V SUM exceeds the integrated feedback voltage (VCOMP).
Synchronous Rectifier Driver (DL)
Synchronous rectification reduces conduction losses in the rectifier by replacing the normal Schottky catch diode with a low-resistance MOSFET switch. The MAX1953/MAX1954/MAX1957 use the synchronous rectifier to ensure proper startup of the boost gatedriver circuit and to provide the current-limit signal. The DL low-side waveform is always the complement of the DH high-side drive waveform. A dead-time circuit monitors the DL output and prevents the high-side MOSFET from turning on until DL is fully off, thus preventing cross-conduction or shoot-through. In order for the dead-time circuit to work properly, there must be a lowresistance, low-inductance path from the DL driver to the MOSFET gate. Otherwise, the sense circuitry in the MAX1953/MAX1954/MAX1957 can interpret the MOSFET gate as OFF when gate charge actually remains. The dead time at the other edge (DH turning off) is determined through gate sensing as well.
Current-Limit Circuit
The current-limit circuit employs a lossless current-limiting algorithm that uses the low-side and high-side MOSFETs' on-resistances as the sensing elements. The voltage across the high-side MOSFET is monitored for current-mode feedback, as well as current limit. This signal is amplified by the current-sense amplifier and is compared with a current-sense voltage. If the currentsense signal is larger than the set current-limit voltage, the high-side MOSFET turns off. Once the high-side MOSFET turns off, the low-side MOSFET is monitored for current limit. If the voltage across the low-side MOSFET (RDS(ON) IINDUCTOR) does not exceed the shortcircuit current limit, the high-side MOSFET turns on normally. In this condition, the output drops smoothly out of regulation. If the voltage across the low-side MOSFET exceeds the short-circuit current-limit threshold at the beginning of each new oscillator cycle, the MAX1953/MAX1954/MAX1957 do not turn on the highside MOSFET. In the case where the output is shorted, the low-side MOSFET is monitored for current limit. The low-side MOSFET is held on to let the current in the inductor ramp down. Once the voltage across the low-side MOSFET drops below the short-circuit current-limit threshold, the high-side MOSFET is pulsed. Under this condition, the frequency of the MAX1953/MAX1954/ MAX1957 appears to decrease because the on-time of the low-side MOSFET extends beyond a clock cycle. The actual peak output current is greater than the short-circuit current-limit threshold by an amount equal to the inductor ripple current. Therefore, the exact current-limit characteristic and maximum load capability are a function of the low-side MOSFET on-resistance, inductor value, input voltage, and output voltage. The short-circuit current-limit threshold is preset for the MAX1954/MAX1957 at 210mV. The MAX1953, however, has three options for the current-limit threshold: connect ILIM to IN for a 320mV threshold, connect ILIM to GND for 105mV, or leave floating for 210mV.
High-Side Gate-Drive Supply (BST)
Gate-drive voltage for the high-side switch is generated by a flying capacitor boost circuit (Figure 5). The capacitor between BST and LX is charged from the VIN supply up to VIN, minus the diode drop while the lowside MOSFET is on. When the low-side MOSFET is switched off, the stored voltage of the capacitor is stacked above LX to provide the necessary turn-on voltage (VGS) for the high-side MOSFET. The controller then closes an internal switch between BST and DH to turn the high-side MOSFET on.
Undervoltage Lockout
If the supply voltage at IN drops below 2.75V, the MAX1953/MAX1954/MAX1957 assume that the supply voltage is too low to make valid decisions, so the UVLO circuitry inhibits switching and forces the DL and DH gate drivers low. After the voltage at IN rises above 2.8V, the controller goes into the startup sequence and resumes normal operation.
Startup
The MAX1953/MAX1954/MAX1957 start switching when the voltage at IN rises above the UVLO threshold. However, the controller is not enabled unless all four of the following conditions are met: * VIN exceeds the 2.8V UVLO threshold. * The internal reference voltage exceeds 92% of its nominal value (VREF > 1 V). * The internal bias circuitry powers up. * The thermal overload limit is not exceeded.
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller
Design Procedures
Setting the Output Voltage
IN BST DH
MAX1953/MAX1954/MAX1957
To set the output voltage for the MAX1953/MAX1954, connect FB to the center of an external resistor-divider connected between the output to GND (Figures 1 and 2). Select R2 between 8k and 24k, and then calculate R1 by: V R1 = R2 x OUT - 1 VFB where VFB = 0.8V. R1 and R2 should be placed as close to the IC as possible.
MAX1953 MAX1954 MAX1957
LX DL
Figure 5. DH Boost Circuit
Once these conditions are met, the step-down controller enables soft-start and starts switching. The soft-start circuitry gradually ramps up to the feedback-regulation voltage in order to control the rate-of-rise of the output voltage and reduce input surge currents during startup. The soft-start period is 1024 clock cycles (1024/fS, MAX1954/MAX1957) or 4096 clock cycles (4096/f S, MAX1953) and the internal soft-start DAC ramps the voltage up in 64 steps. The output reaches regulation when soft-start is completed, regardless of output capacitance and load.
For the MAX1957, connect FB directly to the output through a decoupling resistor of 10k to 21k (Figure 3). The output voltage is then equal to the voltage at REFIN. Again, this resistor should be placed as close to the IC as possible.
Determining the Inductor Value
There are several parameters that must be examined when determining which inductor is to be used. Input voltage, output voltage, load current, switching frequency, and LIR. LIR is the ratio of inductor current ripple to DC load current. A higher LIR value allows for a smaller inductor, but results in higher losses and higher output ripple. A good compromise between size, efficiency, and cost is an LIR of 30%. Once all of the parameters are chosen, the inductor value is determined as follows: L= VOUT x VIN - VOUT
Shutdown
The MAX1953/MAX1954/MAX1957 feature a low-power shutdown mode. Use an open-collector transistor to pull COMP low to shut down the IC. During shutdown, the output is high impedance. Shutdown reduces the quiescent current (IQ) to approximately 220A.
(
VIN x fS x ILOAD MAX x LIR
()
)
Thermal Overload Protection
Thermal overload protection limits total power dissipation in the MAX1953/MAX1954/MAX1957. When the junction temperature exceeds TJ = +160C, an internal thermal sensor shuts down the device, allowing the IC to cool. The thermal sensor turns the IC on again after the junction temperature cools by 15C, resulting in a pulsed output during continuous thermal overload conditions.
where fS is the switching frequency. Choose a standard value close to the calculated value. The exact inductor value is not critical and can be adjusted in order to make trade-offs among size, cost, and efficiency. Lower inductor values minimize size and cost, but they also increase the output ripple and reduce the efficiency due to higher peak currents. By contrast, higher inductor values increase efficiency, but eventually resistive losses due to extra turns of wire exceed the benefit gained from lower AC current levels. For any area-restricted applications, find a low-core loss inductor having the lowest possible DC resistance. Ferrite cores are often the best choice, although powdered iron is inexpensive and can work well at 300kHz.
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15
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
The chosen inductor's saturation current rating must exceed the expected peak inductor current (IPEAK). Determine IPEAK as: LIR IPEAK = ILOAD(MAX ) + x ILOAD(MAX ) 2 FET. A good general rule is to allow 0.5% additional resistance for each C of MOSFET junction temperature rise. The calculated VVALLEY must be less than VCS. For the MAX1953, connect ILIM to GND for a shortcircuit current-limit voltage of 105mV, to VIN for 320mV or leave ILIM floating for 210mV.
Setting the Current Limit
The MAX1953/MAX1954/MAX1957 use a lossless current-sense method for current limiting. The voltage drops across the MOSFETs created by their on-resistances are used to sense the inductor current. Calculate the current-limit threshold as follows: VCS = 0.8V A CS
MOSFET Selection
The MAX1953/MAX1954/MAX1957 drive two external, logic-level, N-channel MOSFETs as the circuit switch elements. The key selection parameters are: * On-Resistance (RDS(ON)): The lower, the better. * Maximum Drain-to-Source Voltage (VDSS): Should be at least 20% higher than the input supply rail at the high side MOSFET's drain. * Gate Charges (Qg, Qgd, Qgs): The lower, the better. For a 3.3V input application, choose a MOSFET with a rated RDS(ON) at VGS = 2.5V. For a 5V input application, choose the MOSFETs with rated RDS(ON) at VGS 4.5V. For a good compromise between efficiency and cost, choose the high-side MOSFET (N1) that has conduction losses equal to switching loss at the nominal input voltage and output current. The selected low-side and highside MOSFETs (N2 and N1, respectively) must have RDS(ON) that satisfies the current-limit setting condition above. For N2, make sure that it does not spuriously turn on due to dV/dt caused by N1 turning on, as this would result in shoot-through current degrading the efficiency. MOSFETs with a lower Qgd/Qgs ratio have higher immunity to dV/dt. For proper thermal management design, the power dissipation must be calculated at the desired maximum operating junction temperature, T J(MAX). N1 and N2 have different loss components due to the circuit operation. N2 operates as a zero-voltage switch; therefore, major losses are the channel conduction loss (PN2CC) and the body diode conduction loss (PN2DC): USE RDS(ON)AT TJ(MAX) V PN2CC = (1 - OUT ) x I2LOAD x RDS(ON) VIN PN2DC = 2 x ILOAD x VF x tDT x fS where VF is the body diode forward-voltage drop, tdt is the dead time between N1 and N2 switching transitions, and fS is the switching frequency.
where ACS is the gain of the current-sense amplifier. ACS is 6.3 for the MAX1953 when ILIM is connected to GND and 3.5 for the MAX1954/MAX1957, and for the MAX1953 when ILIM is connected to IN or floating. The 0.8V is the usable dynamic range of COMP (VCOMP). Initially, the high-side MOSFET is monitored. Once the voltage drop across the high-side MOSFET exceeds VCS, the high-side MOSFET is turned off and the low-side MOSFET is turned on. The voltage across the low-side MOSFET is then monitored. If the voltage across the lowside MOSFET exceeds the short-circuit current limit, a short-circuit condition is determined and the low-side MOSFET is held on. Once the monitored voltage falls below the short-circuit current-limit threshold, the MAX1953/MAX1954/MAX1957 switch normally. The shortcircuit current-limit threshold is fixed at 210mV for the MAX1954/ MAX1957 and is selectable for the MAX1953. When selecting the high-side MOSFET, use the following method to verify that the MOSFET's RDS(ON) is sufficiently low at the operating junction temperature (TJ): RDS(ON)N1 0.8V A CS x IPEAK
The voltage drop across the low-side MOSFET at the valley point and at ILOAD(MAX) is:
LIR VVALLEY = RDS(ON) x (ILOAD(MAX) - x ILOAD(MAX ) ) 2
where RDS(ON) is the maximum value at the desired maximum operating junction temperature of the MOS-
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller
N1 operates as a duty-cycle control switch and has the following major losses: the channel conduction loss (PN1CC), the voltage and current overlapping switching loss (PN1SW), and the drive loss (PN1DR).
V PN1CC = OUT x I2 LOAD x RDS(ON) USE RDS(ON) AT TJ(MAX) VIN Q + QGD PN2SW = VIN x ILOAD x GS x fS IGATE
mended due to their low ESR and ESL at high frequency, with relatively low cost. Choose a capacitor that exhibits less than 10C temperature rise at the maximum operating RMS current for optimum long-term reliability.
MAX1953/MAX1954/MAX1957
(
)
Output Capacitor
The key selection parameters for the output capacitor are the actual capacitance value, the equivalent series resistance (ESR), the equivalent series inductance (ESL), and the voltage-rating requirements. These parameters affect the overall stability, output voltage ripple, and transient response. The output ripple has three components: variations in the charge stored in the output capacitor, the voltage drop across the capacitor's ESR, and the voltage drop across the ESL caused by the current into and out of the capacitor: VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) + VRIPPLE(ESL) The output voltage ripple as a consequence of the ESR, ESL, and output capacitance is: VRIPPLE(ESR) = IP-P x ESR 8 x COUT x fS V VRIPPLE(ESL) = IN ESL L V -V OUT x VOUT IP-P = IN fS x L VIN where IP-P is the peak-to-peak inductor current (see the Determining the Inductor Value section). These equations are suitable for initial capacitor selection, but final values should be chosen based on a prototype or evaluation circuit. As a general rule, a smaller current ripple results in less output voltage ripple. Since the inductor ripple current is a factor of the inductor value and input voltage, the output voltage ripple decreases with larger inductance, and increases with higher input voltages. Ceramic capacitors are recommended for the MAX1953 due to its 1MHz switching frequency. For the MAX1954/ MAX1957, using polymer, tantalum, or aluminum electrolytic capacitors is recommended. The aluminum electrolytic capacitor is the least expensive; however, it has higher ESR. To compensate for this, use a ceramic capacitor in parallel to reduce the switching ripple and noise. For reliable and safe operation, ensure that the capacitor's voltage and ripple-current ratings exceed the calculated values. VRIPPLE(C) IP-P
where IGATE is the average DH driver output current capability determined by: IGATE 1 VIN x 2 RDH + RGATE
where RDH is the high-side MOSFET driver's on-resistance (3 max) and RGATE is the internal gate resistance of the MOSFET (~ 2): PN1DR = QG x VGS x fS x RGATE RGATE + RDH
where VGS ~ VIN. In addition to the losses above, allow about 20% more for additional losses due to MOSFET output capacitances and N2 body diode reverse recovery charge dissipated in N1 that exists, but is not well defined in the MOSFET data sheet. Refer to the MOSFET data sheet for the thermal-resistance specification to calculate the PC board area needed to maintain the desired maximum operating junction temperature with the above calculated power dissipations. The minimum load current must exceed the high-side MOSFET's maximum leakage current over temperature if fault conditions are expected.
Input Capacitor
The input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuit's switching. The input capacitor must meet the ripple current requirement (IRMS) imposed by the switching currents defined by the following equation: IRMS = ILOAD x VOUT x (VIN - VOUT ) VIN
I RMS has a maximum value when the input voltage equals twice the output voltage (VIN = 2 x VOUT), where IRMS(MAX) = ILOAD/2. Ceramic capacitors are recom-
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17
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
The MAX1953/MAX1954/MAX1957s' response to a load transient depends on the selected output capacitors. In general, more low-ESR output capacitance results in better transient response. After a load transient, the output voltage instantly changes by ESR ILOAD. Before the controller can respond, the output voltage deviates further, depending on the inductor and output capacitor values. After a short period of time (see the Typical Operating Characteristics), the controller responds by regulating the output voltage back to its nominal state. The controller response time depends on its closed-loop bandwidth. With a higher bandwidth, the response time is faster, preventing the output voltage from further deviation from its regulating value. Below are equations that define the power modulator: GMOD = gmc x RLOAD x (fS x L) RLOAD + fS x L
(
)
where RLOAD = VOUT/IOUT(MAX), and gmc = 1/(ACS RDS(ON)), where ACS is the gain of the current-sense amplifier and RDS(ON) is the on-resistance of the highside power MOSFET. ACS is 6.3 for the MAX1953 when ILIM is connected to GND, and 3.5 for the MAX1954/ MAX1957 and for the MAX1953 when ILIM is connected to VIN or floating. The frequencies at which the pole and zero due to the power modulator occur are determined as follows:
fpMOD = 1 R LOAD x fS x L + RESR 2 x COUT x RLOAD + fS x L
Compensation Design
The MAX1953/MAX1954/MAX1957 use an internal transconductance error amplifier whose output compensates the control loop. The external inductor, highside MOSFET, output capacitor, compensation resistor, and compensation capacitors determine the loop stability. The inductor and output capacitors are chosen based on performance, size, and cost. Additionally, the compensation resistor and capacitors are selected to optimize control-loop stability. The component values shown in the Typical Application Circuits (Figures 1 through 4) yield stable operation over the given range of input-to-output voltages and load currents. The controller uses a current-mode control scheme that regulates the output voltage by forcing the required current through the external inductor. The MAX1953/ MAX1954/MAX1957 use the voltage across the highside MOSFET's on-resistance (RDS(ON)) to sense the inductor current. Current-mode control eliminates the double pole in the feedback loop caused by the inductor and output capacitor, resulting in a smaller phase shift and requiring less elaborate error-amplifier compensation. A simple single-series RC and CC is all that is needed to have a stable high bandwidth loop in applications where ceramic capacitors are used for output filtering. For other types of capacitors, due to the higher capacitance and ESR, the frequency of the zero created by the capacitance and ESR is lower than the desired close loop crossover frequency. Another compensation capacitor should be added to cancel this ESR zero. The basic regulator loop may be thought of as a power modulator, output feedback divider, and an error amplifier. The power modulator has DC gain set by gmc x RLOAD, with a pole and zero pair set by RLOAD, the output capacitor (COUT), and its equivalent series resistance (RESR).
(
(
)
)
fzMOD =
1 2 x COUT x RESR
The feedback voltage-divider used has a gain of GFB = VFB/VOUT, where VFB is equal to 0.8V. The transconductance error amplifier has DC gain, GEA(DC) = gm RO. RO is typically 10M. A dominant pole is set by the compensation capacitor (C C ), the amplifier output resistance (RO), and the compensation resistor (RC). A zero is set by the compensation resistor (RC) and the compensation capacitor (CC). There is an optional pole set by Cf and RC to cancel the output capacitor ESR zero if it occurs before crossover frequency (fC): 1 2 x CC x (RO + RC ) 1 fzEA = 2 x C C x R C 1 fpEA = 2 x C f x R C fpdEA = The crossover frequency (fC) should be much higher than the power modulator pole f pMOD . Also, the crossover frequency should be less than 1/5 the switching frequency: f fpMOD << fC < S 5
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
Table 2. Suggested Manufacturers
MANUFACTURER Central Semiconductor Coilcraft Fairchild Kemet Panasonic Taiyo Yuden Toko COMPONENT Diode Inductors MOSFETs Capacitors Capacitors Capacitors Inductors PHONE 631-435-1110 800-322-2645 800-341-0392 864-963-6300 714-373-7366 408-573-4150 800-745-8656 WEBSITE www.centralsemi.com www.coilcraft.com www.fairchildsemi.com www.kemet.com www.panasonic.com www.t-yuden.com www.toko.com
so the loop-gain equation at the crossover frequency is: GEA ( fC ) x GMOD( fC ) x VFB =1 VOUT
Applications Information
See Table 2 for suggested manufacturers of the components used with the MAX1953/MAX1954/MAX1957.
PC Board Layout Guidelines
For the case where fzESR is greater than fc: GEA ( fC ) = gmEA x RC and GMOD( fC ) = gmc x fpMOD RLOAD x (fs x L) x RLOAD + (fs x L) fC VOUT gmEA x VFB x GMOD( fC ) Careful PC board layout is critical to achieve low switching losses and clean, stable operation. The switching power stage requires particular attention. Follow these guidelines for good PC board layout: 1) Place decoupling capacitors as close to IC pins as possible. Keep separate power ground plane (connected to pin 7) and signal ground plane (connected to pin 4). 2) Input and output capacitors are connected to the power ground plane; all other capacitors are connected to the signal ground plane. 3) Keep the high current paths as short as possible. 4) Connect the drain leads of the power MOSFET to a large copper area to help cool the device. Refer to the power MOSFET data sheet for recommended copper area. 5) Ensure all feedback connections are short and direct. Place the feedback resistors as close to the IC as possible. 6) Route high-speed switching nodes away from sensitive analog areas (FB, COMP). 7) Place the high-side MOSFET as close as possible to the controller and connect IN (MAX1953/MAX1957) or HSD (MAX1954) and LX to the MOSFET. 8) Use very short, wide traces (50mils to 100mils wide if the MOSFET is 1in from the device).
then RC is calculated as: RC =
where gmEA = 110S. The error amplifier compensation zero formed by RC and CC should be set at the modulator pole fpMOD. CC is calculated by: VOUT CC = IOUT(MAX) VOUT IOUT(MAX) x (fS x L) x + (fS x L) COUT RC
As the load current decreases, the modulator pole also decreases. However, the modulator gain increases accordingly, and the crossover frequency remains the same. For the case where fzESR is less than fC, add another compensation capacitor Cf from COMP to GND to cancel the ESR zero at fzESR. Cf is calculated by: 1 Cf = 2 x RC x fzESR Figure 6 illustrates a numerical example that calculates RC and CC values for the typical application circuit of Figure 1 (MAX1953).
Chip Information
TRANSISTOR COUNT: 2930 PROCESS: BiCMOS
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19
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
VOUT = 2.5V IOUT(MAX) = 3A COUT = 20F L = 1H RESR = 0.0025 gmEA = 110S A VCS = 6.3A RDS(ON) = 0.013 1 = 12.21S A VCS x RDS(ON) fS = 1MHz VOUT 2.5V RLOAD = = = 0.833 IOUT(MAX) 3A gmc = fpMOD = 1 R LOAD x fS x L 2 x COUT x R LOAD fS x L
(
(
)
)
+ RESR )
=
1 = 17.42kHz 0.833 x 1MHz x 1H + 0.0025 2 x 20F x 0.833 + 1MHz x 1H
(
)
= 3.2MHz 2 x COUT x RESR 2 x 20F x .0025 Pick the crossover frequency (fC ) at < 1/ 5 the switching frequency (fS ). We choose 100kHz < fzESR, so CF is not needed. The power modulator gain at fC is : GMOD(fC ) = gmc x then : RC = And : VOUT CC = IOUT(MAX) VOUT IOUT(MAX) x (fS x L) x + (fS x L) 2.5V COUT RC = 3A 2.5V 3A x (1MHz x 1H) + (1MHz x 1H) 20F 33k VOUT gmEA x VFB x GMOD(fC ) = 2.5V 110S x 0.8V x .937 33k fpMOD 0.833 x (1MHz x 1H) RLOAD x (fS x L) 17.42kHz x = 0.967 x = 12.21S x 0.833 + (1MHz x 1H) 100kHz RLOAD (fS x L) fC
fzESR =
1
=
1
x
270pF
Figure 6. Numerical Example to Calculate RC and CC Values of the Typical Operating Circuit of Figure 1 (MAX1953)
20
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Low-Cost, High-Frequency, Current-Mode PWM Buck Controller
Pin Configurations (continued)
MAX1953/MAX1954/MAX1957
TOP VIEW
HSD 1 COMP FB GND IN 2 3 4 5 10 BST 9 LX DH PGND DL REFIN 1 COMP FB GND IN 2 3 4 5 10 BST 9 LX DH PGND DL
MAX1954EUB
8 7 6
MAX1957EUB
8 7 6
MAX
MAX
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21
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.)
10LUMAX.EPS
1 1
e
10
4X S
10
INCHES MAX DIM MIN A 0.043 A1 0.006 0.002 A2 0.030 0.037 D1 0.116 0.120 D2 0.114 0.118 E1 0.116 0.120 E2 0.114 0.118 H 0.187 0.199 L 0.0157 0.0275 L1 0.037 REF b 0.007 0.0106 e 0.0197 BSC c 0.0035 0.0078 0.0196 REF S 0 6
MILLIMETERS MAX MIN 1.10 0.15 0.05 0.75 0.95 3.05 2.95 2.89 3.00 2.95 3.05 2.89 3.00 4.75 5.05 0.40 0.70 0.940 REF 0.177 0.270 0.500 BSC 0.090 0.200 0.498 REF 0 6
H y 0.500.1 0.60.1
1
1
0.60.1
TOP VIEW
BOTTOM VIEW
D2 GAGE PLANE A2 A b D1 A1
E2 c E1 L1
L
FRONT VIEW
SIDE VIEW
PROPRIETARY INFORMATION TITLE:
PACKAGE OUTLINE, 10L uMAX/uSOP
APPROVAL DOCUMENT CONTROL NO. REV.
21-0061
I
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
22 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2002 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

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